1. Field of the Invention
The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.
2. Description of the Related Art
Most of power supply circuits that rectify commercial alternating-current power and provide desired direct-current voltage have recently been switching type power supply circuits. A switching power supply circuit has a transformer and other devices miniaturized by increasing switching frequency, and is used as a power supply for various electronic devices as a high-power DC-to-DC converter.
The commercial alternating-current power is a sinusoidal alternating voltage. When a smoothing and rectifying circuit using a rectifying element and a smoothing capacitor rectifies and smoothes the commercial alternating-current power, due to a peak hold effect of the smoothing and rectifying circuit, a current flows from the commercial alternating-current power supply to the switching power supply circuit only during a short period around a peak voltage of the alternating voltage, and the current flowing from the commercial alternating-current power supply to the switching power supply circuit has a distorted waveform that differs greatly from a sinusoidal wave. Then a power factor indicating efficiency of use of the power supply is deteriorated. In addition, a measure is required to suppress harmonics of the cycle of the commercial alternating-current power which harmonics result from such a distorted current waveform. A method using a so-called active filter to solve these problems is known as a conventional technology for improving the power factor (for example Japanese Patent Laid-Open No. Hei 6-327246).
FIG. 24 shows a basic configuration of such an active filter. In FIG. 24, a primary side rectifying element Di formed as a bridge rectifier is connected to a commercial alternating-current power supply AC. A step-up type converter is connected to the positive electrode/negative electrode line of the primary side rectifying element Di. A smoothing capacitor Cout is connected in parallel with the output of the converter. A direct-current voltage Vout is obtained as voltage across the smoothing capacitor Cout. The direct-current voltage Vout is supplied as input voltage to a load 110 such for example as a DC-to-DC converter in a subsequent stage.
A configuration for power factor improvement includes: the step-up type converter composed of an inductor L, a fast recovery type fast switching diode D, and a switching element Q; and a control section for the step-up type converter, the control section having a multiplier 111 as a main component. The inductor L and the fast switching diode D are inserted between the positive electrode output terminal of the primary side rectifying element Di and the positive electrode terminal of the smoothing capacitor Cout in a state of being connected in series with each other. A resistance Ri is inserted between the negative electrode output terminal of the primary side rectifying element Di (primary side ground) and the negative electrode terminal of the smoothing capacitor Cout. The switching element Q is a MOS-FET, for example. The switching element Q is inserted between a point of connection between the inductor L and the fast switching diode D and the primary side ground.
The multiplier 111 is connected with a current detection line LI, a waveform input line Lw, and a voltage detection line Lv. The multiplier 111 detects a signal corresponding to a rectified current Iin flowing through the negative electrode output terminal of the primary side rectifying element Di from across the resistance Ri, the signal being input from the current detection line LI. In addition, the multiplier 111 detects a signal corresponding to a rectified voltage Vin at the positive electrode output terminal of the primary side rectifying element Di, the signal being input from the waveform input line Lw. This rectified voltage Vin is obtained by converting the waveform of an alternating input voltage VAC from the commercial alternating-current power supply AC into an absolute value. Further, the multiplier 111 detects a variation difference of the direct-current input voltage (a signal obtained by amplifying a difference between a predetermined reference voltage and the direct-current voltage Vout will be referred to as a variation difference, which will hereinafter be used similarly) on the basis of the direct-current voltage Vout of the smoothing capacitor Cout, the direct-current voltage Vout being input from the voltage detection line Lv. Then, a drive signal for driving the switching element Q is output from the multiplier 111.
The multiplier 111 multiplies together the signal corresponding to the rectified current Iin, the signal being detected from the current detection line LI, and the variation difference of the direct-current input voltage, the variation difference being detected from the voltage detection line Lv. The multiplier 111 detects an error between a result of the multiplication and the signal corresponding to the rectified voltage Vin, the signal being detected from the waveform input line Lw. After amplifying the error signal, the multiplier 111 performs a PWM (Pulse Width Modulation) conversion, and controls the switching element Q by a binary signal having a high level and a low level. Thus, a two-input feedback system is formed, the value of the direct-current voltage Vout is made to be a predetermined value, and the rectified current Iin is made to have a waveform similar to that of the rectified voltage Vin. As a result, the waveform of the alternating voltage applied from the commercial alternating-current power supply AC to the primary side rectifying element Di and the waveform of the alternating current flowing into the primary side rectifying element Di are also similar to each other, so that the power factor approaches substantially one. Thus power factor improvement is achieved.
FIG. 25A shows the rectified voltage Vin and the rectified current Iin when the active filter circuit shown in FIG. 24 operates properly. FIG. 25B shows change Pchg in energy (power) input and output to and from the smoothing capacitor Cout. A broken line represents an average value Pin of the input and output energy (power). That is, the smoothing capacitor Cout stores energy when the rectified voltage Vin is high, and the smoothing capacitor Cout releases energy when the rectified voltage Vin is low. The smoothing capacitor Cout thereby maintains a flow of output power. FIG. 25C shows the waveform of a charging and discharging current Ichg of the smoothing capacitor Cout. FIG. 25D shows the direct-current voltage Vout as voltage across the smoothing capacitor Cout. The direct-current voltage Vout is a direct-current voltage (for example a direct-current voltage of 375 V) on which a ripple voltage including a second harmonic component of the cycle of the rectified voltage Vin as a main component is superimposed.
FIG. 26 shows an example of configuration of a power supply circuit formed by connecting a current resonant converter in a stage following an active filter based on the configuration shown in FIG. 24. The power supply circuit shown in FIG. 26 has a configuration that can deal with load power Po in a range of 300 W to 0 W when the value of the alternating input voltage VAC is in a range of 85 V to 264 V. The current resonant converter employs the configuration of an externally excited half-bridge coupling system.
The power supply circuit shown in FIG. 26 will be described in order from an alternating current input side. A common mode noise filter formed by two line filter transformers LFT and three across capacitors CL is provided. A primary side rectifying element Di is connected in a stage subsequent to the common mode noise filter. A pi-configuration normal mode noise filter 125 formed by connecting an inductor LN and filter capacitors (film capacitors) CN is connected to the rectified output line of the primary side rectifying element Di.
The positive electrode output terminal of the primary side rectifying element Di is connected to the positive electrode terminal of a smoothing capacitor Ci via a series connection of the inductor LN, a choke coil PCC (functioning as an inductor Lpc), and a fast recovery type fast switching diode D20. The smoothing capacitor Ci has the same function as the smoothing capacitor Cout in FIG. 24. The inductor Lpc of the choke coil PCC and the fast switching diode D20 have the same function as the inductor L and the fast switching diode D, respectively, shown in FIG. 24. In addition, an RC snubber circuit formed by a capacitor Csn and a resistance Rsn connected in series with each other is connected in parallel with the fast switching diode D20 in FIG. 26.
A switching element Q103 corresponds to the switching element Q in FIG. 24. A power factor and output voltage controlling IC 120 in this case is an integrated circuit (IC) that controls operation of the active filter for improving a power factor so as to approximate the power factor to one. The power factor and output voltage controlling IC 120 includes for example a multiplier, a divider, an error voltage amplifier, a PWM control circuit, and a drive circuit for outputting a drive signal for driving the switching element Q103. A first feedback control circuit for setting a direct-current input voltage Ei at a predetermined value is formed by inputting a voltage obtained by dividing a voltage across the smoothing capacitor Ci (direct-current input voltage Ei) by a voltage dividing resistance R5 and a voltage dividing resistance R6 to a terminal T1 of the power factor and output voltage controlling IC 120.
In addition, a series connection of a voltage dividing resistance R101 and a voltage dividing resistance R102 is provided between the positive electrode output terminal of the primary side rectifying element Di and a primary side ground. A point of connection between the voltage dividing resistance R101 and the voltage dividing resistance R102 is connected to a terminal T5. Thereby a voltage rectified by the primary side rectifying element Di is divided and then input to the terminal T5. The voltage of a resistance 103, that is, a voltage corresponding to the source current of the switching element Q103 is input to a terminal T2. The source current of the switching element Q103 is a current contributing to storing of magnetic energy, in a current I101 flowing through the choke coil PCC. Then, a second feedback control circuit is formed which makes the signal corresponding to the rectified voltage which signal is input to the terminal T5 of the power factor and output voltage controlling IC 120 have a similar form to that of the signal corresponding to the envelope of the voltage input to the terminal T2 (that is, the envelope of the current I101).
In addition, a terminal T4 is supplied with operating power for the power factor and output voltage controlling IC 120. A half-wave rectifier circuit formed by a rectifier diode D11 and a series resonant capacitor C11 shown in FIG. 26 converts an alternating voltage induced in a winding N5, which is transformer-coupled with the inductor Lpc in the choke coil PCC, into a low direct-current voltage, and then supplies the low direct-current voltage to the terminal T4. In addition, the terminal T4 is connected to the positive electrode output terminal of the primary side rectifying element Di via a starting resistance Rs. During a start-up time before a voltage is induced in the winding N5 after turning on the commercial alternating-current power supply AC, the rectified output obtained at the positive electrode output terminal of the primary side rectifying element Di is supplied to the terminal T4 via the starting resistance Rs. The power factor and output voltage controlling IC 120 starts operation using the thus supplied rectified voltage as starting power.
A drive signal (gate voltage) for driving the switching element is output from a terminal T3 to the gate of the switching element Q103. That is, the drive signal for operating the two feedback control circuits, that is, the first feedback control circuit for making the value of the voltage divided by the above-mentioned voltage dividing resistances R5 and R6 a predetermined value and the second feedback control circuit for making the envelope of the current I101 have a similar form to that of the direct-current input voltage Ei is input to the gate of the switching element Q103. Thereby the waveform of an alternating input current IAC flowing in from the commercial alternating-current power supply AC is substantially the same as the waveform of the alternating input voltage VAC, so that the power factor is controlled to be substantially one. That is, the power factor is improved.
FIGS. 27A, 27B, and 27C and FIG. 28 show the waveforms of parts in power factor improving operation of the active filter shown in FIG. 26. FIGS. 27A, 27B, and 27C show the switching operation (on: conducting, and off: disconnecting operation) of the switching element Q103 and the current I101 flowing through the inductor Lpc of the choke coil PCC according to load variation. FIG. 27A shows operation under a light load. FIG. 27B shows operation under a medium load. FIG. 27C shows operation under a heavy load. As is understood from comparison between FIG. 27A, FIG. 27B, and FIG. 27C, the switching cycle of the switching element Q103 is held constant, while the on period of the switching element Q103 is lengthened as the load becomes heavier. The current I101 flowing into the smoothing capacitor Ci via the inductor Lpc is thus adjusted according to the load condition, whereby the direct-current input voltage Ei is stabilized irrespective of voltage variation of the alternating input voltage VAC and load variation. For example, the value of the direct-current input voltage Ei is held constant at 380 V while the value of the alternating input voltage VAC is in a range of 85 V to 264 V. The direct-current input voltage Ei is a voltage across the smoothing capacitor Ci, and is a direct-current input voltage for the current resonant converter in the following stage.
FIG. 28 shows the waveforms of the alternating input current IAC and the direct-current input voltage Ei for comparison with the alternating input voltage VAC. Incidentally, this figure shows results of an experiment when the value of the alternating input voltage VAC is 100 V. As shown in this figure, the waveform of the alternating input voltage VAC and the waveform of the alternating input current IAC are substantially similar to each other with the passage of time. That is, the power factor is improved. In addition to such an improvement in power factor, it is shown that the direct-current input voltage Ei is stabilized at an average value of 380 V. Also, as shown in FIG. 28, the direct-current input voltage Ei has ripple variations of 10 Vp-p at 380 V.
Returning to FIG. 26, description will be made of the current resonant converter in the stage following the active filter. The current resonant converter is supplied with the direct-current input voltage Ei and performs switching operation for power conversion. The current resonant converter has a switching circuit formed with switching elements Q101 and Q102 connected by a half-bridge connection. The current resonant converter in this case is externally excited. MOS-FETs are used as the switching element Q101 and the switching element Q102. A body diode DD101 and a body diode DD102 are respectively connected in parallel with these MOS-FETS. An oscillating and driving circuit 102 switching-drives the switching element Q101 and the switching element Q102 at a required switching frequency in timing in which the switching element Q101 and the switching element Q102 are alternately turned on/off. The oscillating and driving circuit 102 is controlled by a signal from a control circuit 101. The control circuit 101 operates so as to variably control the switching frequency according to the level of a secondary side direct-current output voltage Eo. Thereby the secondary side direct-current output voltage Eo is stabilized.
A converter transformer PIT is provided to transmit the switching output of the switching element Q101 and the switching element Q102 from a primary side to a secondary side. One terminal part of a primary winding N1 of the converter transformer PIT is connected to a point of connection between the switching element Q101 and the switching element Q102 (switching output point) via a primary side series resonant capacitor C101. Another terminal part of the primary winding N1 is connected to the primary side ground. The primary side series resonant capacitor C101 and a primary side leakage inductance L1 form a series resonant circuit. The series resonant circuit performs a resonant operation by being supplied with the switching output by the switching element Q101 and the switching element Q102.
A secondary winding N2 is wound on the secondary side of the converter transformer PIT. The secondary winding N2 in this case has a secondary winding part N2A and a secondary winding part N2B provided with a center tap as shown in FIG. 26. The center tap is connected to a secondary side ground. The secondary winding part N2A and the secondary winding part N2B are connected to the anodes of a rectifier diode Do1 and a rectifier diode Do2, respectively. The cathodes of the rectifier diode Do1 and the rectifier diode Do2 are each connected to a smoothing capacitor Co. Thus a double-wave rectifier circuit is formed. Thereby the secondary side direct-current output voltage Eo is obtained as voltage across the smoothing capacitor Co. This secondary side direct-current output voltage Eo is supplied to a load side not shown in the figure and also input to the above-mentioned control circuit 101.
FIG. 29 shows characteristics of power conversion efficiency ηAC→DC (overall efficiency) from AC power to DC power, the power factor PF, and the direct-current input voltage Ei with respect to load variation. FIG. 29 shows the characteristics when the value of the alternating input voltage VAC is 100 V and the value of load power Po is varied from 300 W to 0 W. FIG. 30 shows characteristics of the power conversion efficiency ηAC→DC (overall efficiency), the power factor PF, and the direct-current input voltage Ei with respect to variation in the alternating input voltage VAC. FIG. 30 shows the characteristics when the value of the alternating input voltage VAC is varied from 85 V to 264 V under a load condition where the value of the load power Po is constant at 300 W.
First, as shown in FIG. 29, the power conversion efficiency (overall efficiency) is decreased as the load power Po is increased. With respect to variation in the alternating input voltage VAC, as shown in FIG. 30, the power conversion efficiency (overall efficiency) is increased as the level of the alternating input voltage VAC becomes higher under the same load condition. For example, results obtained show that under the load condition where the load power Po is 300 W, the power conversion efficiency (overall efficiency) is about 83.0% when the alternating input voltage VAC is 100 V, the power conversion efficiency (overall efficiency) is about 89.0% when the alternating input voltage VAC is 230 V, and the power conversion efficiency (overall efficiency) is about 80.0% when the alternating input voltage VAC is 85 V.
The power factor PF is substantially constant as the load power Po is varied, as shown in FIG. 29. As for characteristics of variation of the power factor PF in relation to variation in the alternating input voltage VAC, FIG. 30 shows that although the power factor PF is decreased as the alternating input voltage VAC is increased, the power factor PF may be considered to be substantially constant. For example, under the load condition where the load power Po is 300 W, the value of the power factor PF is about 0.96 when the alternating input voltage VAC is 100 V, and the value of the power factor PF is about 0.94 when the alternating input voltage VAC is 230 V.
As shown in FIG. 29 and FIG. 30, results obtained show that the direct-current input voltage Ei is substantially constant as the load power Po or the alternating input voltage VAC is varied.